The present invention relates generally to vestigial sideband generators or modulators and more particularly to a new and improved vestigial sideband generator or modulator employing modified Weaver modulator techniques and arrangements. Another aspect of the invention relates generally to digital television transmitters and more particularly to a digital television transmitter including a digital modulator including a modified Weaver modulator arrangement and technique.
A typical prior art digital television transmitter adapted to transmit signals containing information indicative of digitally encoded video and aural signals for deriving an ATSC A/53 standard signal is illustrated in FIG. 1 as including multi-bit digital baseband television signal source 10 which drives the cascaded combination of data randomizer 11, Reed-Solomon encoder 12, data interleaver 14, trellis encoder 16 and multiplexer 18. The signals derived from source 10, randomizer 11, encoders 12 and 16, as well as interleaver 14 and multiplexer 18 are typically three or four parallel bit signals having a symbol rate (i.e., sampling frequency) of             1.539      xc3x97              10        9              143    ,
i.e., the encoded television signal is sampled 10,762,237.76 times per second. Because each symbol includes two, three or four bits, the bit rate is substantially higher than the symbol rate. The three or four parallel bits represent 8 or 16 amplitude levels of the encoded television signal.
Multiplexer 18, in addition to being responsive to the output of trellis encoder 18, responds to segment synchronizing source 20 and field synchronizing source 22 to derive an output having the same number of bits as applied to the multiplexer by encoder 16. Multiplexer 14 supplies a multi-bit output signal to pilot inserter 24 which inserts a 309.44056 KHz pilot carrier on the signal applied to it. Pilot inserter 24 derives a multi-bit output signal which it applies to pre-equalizer filter 26. Pre-equalizer filter 26 supplies a multi-bit intermediate frequency (I.F.) signal to vestigial sideband modulator or generator 28. Generator 28 feeds a multi-bit digital I.F. output signal to digital to analog converter 30, which supplies an analog I.F. signal to frequency up converter 32, a frequency synthesizer for heterodyning the I.F. output frequency of converter 30 to a radio frequency (R.F.) carrier frequency. Up converter 32 also inverts the I.F. spectrum derived from digital to analog converter 30 so the lowest frequencies in the I.F. spectrum are converted into the highest frequencies in the R.F. spectrum derived by converter 32 and the highest frequencies in the I.F. spectrum are converted to the lowest frequencies in the R.F. spectrum. The modulated carrier frequency signal derived by R.F. up converter 32 is applied to antenna 34 via power amplifier 36.
The output signal of digital to analog converter 30 includes orthogonal I and Q channels or components. At predetermined time intervals the I channel has one of multiple levels, corresponding to the number of amplitude levels in the 3 or 4 bit signal derived by signal source 10. The Q channel contains no independent information, but causes part of the unwanted lower sideband appearing at the output of up converter 32 to be reduced substantially to zero amplitude. The unwanted lower sideband is removed by circuitry included in vestigial sideband generator 28 and up converter 32 does not reintroduce it. Because up converter 32 xe2x80x9cflipsxe2x80x9d (i.e., inverts) the I.F. spectrum derived by digital to analog converter 30, the upper sideband R.F. output of converter 30 is reduced substantially to zero.
To enable digital to analog converter 30 to produce the desired vestigial sideband signal, vestigial sideband modulator or generator 28 derives the spectrum illustrated in FIG. 2 having a 6 Mhz bandwidth and including the 309.44056 kHz pilot carrier provided by pilot inserter 24, as well as a vestigial sideband of 309.4405594 kHz, to the left of the pilot carrier frequency.
The prior art vestigial sideband modulators or generators for deriving the ATSC A/53 standard have generally used a filter or phasing method. In the filter method the vestigial sideband modulator generates a double sideband signal that is filtered to produce a vestigial sideband signal at an I.F. of about 10 MHz. Sidebands extend equally around the 10 MHz I.F. in accordance with:
0.5Fsym+Fpilot=6 MHzxe2x88x92Fpilot
where Fsym is the symbol clock frequency of 10.76223776 . . . MHz of the bits derived from source 10 in accordance with the ATSC A/53 standard, and       F    pilot    =                    59        3            ·              FH        NTSC              =                  59        3            ·                        4.5          ⁢                      xe2x80x83                    ⁢          MHz                286            
where FHNTSC is the NTSC horizontal line frequency
Based on the foregoing, the sidebands of the double sideband modulator extend xc2x15.690559441 . . . MHz on either side, of the 10 MHz carrier. A convenient sampling frequency is four times the 10.76223776 . . . MHz symbol clock rate, i.e., 43.04895105 . . . MHz.
The ATSC A/53 standard requires the vestigial sideband generator to have a root-raised cosine (RRC) response. Obtaining a proper root-raised cosine response for vestigial sideband shaping at the 43.04895105 . . . MHz sampling rate requires a finite impulse response (FIR) filter having about 2048 filter coefficients. Implementation of such a filter is difficult.
The phasing method uses a Hilbert transform to partially cancel the unwanted sideband of a double sideband signal. The Hilbert transform can easily generate a vestigial sideband signal such that DC is 6 dB down with respect to the sidebands. This is because the response of any Hilbert transform approximation is always zero at DC. With only one of the I and Q modulators included in such a vestigial Hilbert transform sideband generator contributing at DC, the vector sum of the outputs of the two modulators drops in half at DC relative to the vector sum at a frequency where both the I and Q channels contribute to the generator output. However, in the ATSC A/53 standard, the requirement for the root-raised cosine response places the DC output at xe2x88x923 dB instead of xe2x88x926 dB. Therefore, the Hilbert transform method of vestigial sideband digital television modulation requires a low frequency equalizer to produce a +3 dB xe2x80x9cshelfxe2x80x9d at the DC and low frequency portions of the response.
To achieve the ATSC A/53 standard the vestigial sideband generator has a linear phase requirement. Consequently, equalizer filter 26 is generally implemented as a finite impulse response filter having a large number of coefficients. Further, the xe2x88x923 dB requirement exists at the Nyquist frequency of the symbols, i.e., half the symbol frequency, with certain modifications. Hence, equalizer 26 must include a high frequency portion operating at a sampling frequency higher than twice the symbol rate to avoid aliasing, i.e. insertion of information at frequencies that do not exist in the sample frequency due to sampling at a frequency less than twice the highest frequency component being filtered. In this case, the highest frequency being filtered is 5.690559441 . . . MHz, which is more than half the symbol rate. Hence, the Hilbert transform method of producing a vestigial sideband signal with root-raised cosine sideband shaping is also quite difficult to implement.
Because power amplifier 34 has a non-linear amplitude response, a nonlinear equalizer must apply a substantial non-linear correction to the signal applied to it. Because of the possibility of aliasing and spectral folding through zero frequency, the amount of nonlinear correction which may be applied at 10 MHz is limited, resulting in distortion in the transmitted signal.
I have realized that the non-linear correction can be more effectively implemented to substantially reduce distortion in the transmitted signal by employing an I.F. digital signal having a frequency approximately twice the approximately 10 MHz frequency of the prior art digital I.F. The arrangement I have invented enables the digital I..F. signal to have a frequency of approximately 21.5 MHz; I have also devised an arrangement enabling the digital I.F. to be exactly 21.5 MHz.
I have realized that the problems of the prior art vestigial sideband modulators or generators employed in digital television transmitters can be resolved by using digital signal processing techniques similar to analog signal processing techniques used in Weaver single sideband analog modulators; the Weaver single sideband modulator must be modified to enable the vestigial sideband signal to be derived. The digital signal processing techniques I have developed are applicable to digital signals derived in accordance with the ATSC standard A/53, as well as digital television signals derived in accordance with the NTSC SMPTE standard.
A conventional analog Weaver single sideband modulator is illustrated in FIG. 3 as including identical balanced modulators (i.e., mixers) 30 and 32, driven in parallel by analog signal source 34, having any angular frequency xcfx89m in a predetermined bandwidth xcfx891 to xcfx892. Mixers 30 and 32 are driven by orthogonally phased cosine and sine analog waves derived from folding frequency oscillator 36, having an angular frequency xcfx89f. The a frequency, xcfx89f, of the sinusoidal waves derived by oscillator 36 is approximately equal to the arithmetical mean of the bandwidth of the signal derived from source 34, i.e.,                     ω        1            +              ω        2              2    .
Mixers 30 and 32 thereby derive xe2x80x9cfoldedxe2x80x9d baseband signals, each having a bandwidth of approximately one-half the bandwidth of the signal derived from source 34.
The folded baseband signals derived by mixers 30 and 32 are respectively supplied to identical lowpass filters 38 and 40. Filters 38 and 40 have a cut-off frequency designed to (1) virtually completely attenuate (i.e., reject) the upper sidebands derived from mixers 30 and 32, and (2) pass with virtually no attenuation at least half the bandwidth of signal source 34. Filters 38 and 40 have a transition frequency range (i.e., the frequency range over which the filter response changes from substantially maximum to substantially minimum attenuation) no greater than twice the lowest frequency of source 34.
The folded baseband signals derived by lowpass filters 38 and 40 are applied to balanced modulators (i.e., mixers) 42 and 44, respectively driven by orthogonally phased cosine and sine waves at an I.F. or R.F. carrier frequency, derived by oscillator 46. Because mixers 30, 32, 42 and 44 are balanced modulators they produce the upper and lower sidebands of the waves applied to them without passing the I.F. or R.F. of oscillator 46.
The output signals of mixers 42 and 44 are linearly combined (i.e., summed or subtracted) in analog adder 48. If the signal of source 34 is represented by sin xcfx89mt and the angular frequencies of oscillators 36 and 46 are respectively xcfx89f and xcfx89c, it can be shown that the output signal of analog adder 48 is 0.5 sin(xcfx89xe2x88x92xcfx89f+xcfx89c)t when adder 48 sums the output signals of balanced modulators 42 and 44; if adder 48 subtracts the output signal of mixer 44 from the output signal of mixer 42, the output signal of adder 48 is 0.5 sin(xcfx89fxe2x88x92xcfx89m+xcfx89c)t. Thus, when adder 48 sums the output signals of mixers 42 and 44, the adder output signal is the upper sideband of the combination of the frequencies derived by generator 46 and the folded baseband signals derived by filters 38 and 40; the output of adder 48 is the lower sideband when adder 48 is configured to subtract the output of mixer 44 from the output signal of mixer 42 are critical.
While the Weaver modulator theoretically derives a single sideband output signal that is a replica of the baseband signal of FIG. 3a, it has not been extensively employed for processing analog signals because of the need for lowpass filters 38 and 40 to be sharp cut off filters which are accurately matched to have the same amplitude and phase responses. In addition, all the elements of the two parallel paths must be matched and the quadrature phase relationships of xcfx89f and xcfx89c are critical.
To provide a better understanding of how the Weaver modulator of FIG. 3 functions, reference is made to the amplitude versus frequency spectra plots of FIGS. 4a-4e. In FIGS. 4a and 4b, the baseband output signal of source 34 is represented by identical spectra 52 and 54. Each of spectra 52 and 54 has a passband between xcfx891 and xcfx892 relative to DC, i.e., xcfx89=0. Spectra 52 and 54 are respectively multiplied in mixers 30 and 32 by cos xcfx89ft and sin xcfx89ft outputs of oscillator 34, respectively represented by lines 56 and 58 in FIGS. 4a and 4b to produce folded orthogonal baseband components I and Q. The positive frequencies in spectra 60 and 62 at the outputs of lowpass filters 38 and 40 are derived from the portion of spectra 52 and 54 between xcfx89f and xcfx892 while the positive frequencies in spectra 64 and 66 at the outputs of filters 38 and 40 are derived from the portion of spectra 52 and 54 between xcfx891 and xcfx89f. Thus, the baseband spectra 52 and 54 are translated into spectra 60-66, centered on and symmetrical with DC i.e., xcfx89=0; spectra 60-66 extend from   -            (                        ω          1                +                  ω          2                    )        2  
to       +                  (                              ω            1                    +                      ω            2                          )            2        ,
thus include xe2x80x9cnegativexe2x80x9d frequencies resulting from e multiplying action of spectra 52 and 54 with cos xcfx89ft and sin xcfx89ft.
The action of oscillator 46 and mixers 42 and 44 is illustrated in FIGS. 4c and 4d. The folded baseband I and Q spectra 60-66 in the right portion of FIGS. 4a and 4b are shown on the left sides of FIGS. 4c and 4d, and multiplied in mixers 42 and 44 responsive to the cosine and sine outputs at xcfx89c of oscillator 46, as represented by lines 65 and 67, FIGS. 4c and 4d. The resulting outputs of mixers 42 and 44 are represented by spectra 68-74, FIGS. 4c and 4d. All of spectra 68-78 are symmetrical with carrier frequency xcfx89c. Spectra 68 and 70 are respectively replicas of the portions of I and Q spectra 60 and 72 above xcfx89f, while spectra 72 and 74 are respectively replicas of the portions of I and Q spectra 64 and 66 lower than xcfx89f.
FIG. 4e indicates the action of adder 48 in summing the output signals of mixers 42 and 44. Folded I and Q spectra 68 and 70 are summed by adder 48, which derives single upper sideband spectrum 76; all of spectrum 76 lies above xcfx89cxe2x88x92xcfx89f, where xcfx89cxe2x88x92xcfx89f represents the suppressed carrier. The phases of folded I and Q spectra 72 and 74 are such that there is no lower sideband (i.e., frequency inverted) energy. If adder 48 subtracts the output of mixer 44 from the output of mixer 42, the resulting single lower sideband spectrum would be the mirror image of response 76.
I have realized that the Weaver modulation method can produce vestigial sideband signals by selecting the folding frequency of oscillator 36 and the cutoff frequencies of filters 38 and 40 such that the xe2x80x9cnegativexe2x80x9d frequencies of the modulating baseband signal derived from source 34 appear in the signal derived from adder 48 as a vestige of the opposite sideband, i.e., the positive frequencies of the modulating baseband signal. The folding frequency of oscillator 36 and the cutoff frequency of lowpass filters 38 and 40 are selected such that a desired amount of the opposite sideband is eliminated but a certain portion of it is passed. Hence, the Weaver modulator circuit configuration to achieve a vestigial sideband signal is the same as illustrated in FIG. 3, except for modifications of the frequency of oscillator 36 and the cutoff frequency of filters 38 and 40. Preferably, vestigial sideband modulation is produced by modifying the Weaver modulator so oscillator 36 has a reduced frequency and filters 38 and 40 have increased cutoff frequencies. Digital signal processing techniques easily establish the matched paths, which are difficult or impossible to achieve in analog processing.
While the folding frequency is preferably equal to or less than the center of the spectrum of the input signal, i.e.,             ω      f        ≤                            ω          1                +                  ω          2                    2        ,
(where xcfx891 and xcfx892 are respectively the lower and upper angular frequencies of the spectrum of the input signal), there are no theoretical limits to the folding frequency, except that it cannot be zero. A folding frequency of zero would not produce any sideband asymmetry. However, if the folding frequency has any non-zero value the modulator derives vestigial sideband signals with different vestigial widths. As long as the folding frequency is somewhere within the bandwidth of the input signal the intermediate I and Q channel spectra will fold.
However, if the folding frequency is higher than the highest modulating frequency, vestigial modulation will result but there will be no xe2x80x9cfoldingxe2x80x9d through zero frequency. For example, if the signal bandwidth is DC to 3 kHz, and the xe2x80x9cfoldingxe2x80x9d frequency is 4 kHz, the I and Q signals which are not actually folded extend from 1 to 7 kHz. The modulator produces a vestigial sideband signal derived in this instance by lowpass filtering the product of the wave at the xe2x80x9cfoldingxe2x80x9d frequency and the input signal somewhere between 4 and 7 kHz. There is normally no advantage to locating the folding frequency outside the bandwidth of the modulating input signal. Technical disadvantages of locating the folding frequency outside the bandwidth of the input modulating input signal include requirements for higher sampling rates and more complicated filters than occurs for folding frequencies within the bandwidth of the input modulating input signal. While the spectra will not xe2x80x9cfoldxe2x80x9d if the folding frequency is outside the signal bandwidth the modulator could still work if higher sampling rates and more complicated filters are employed.
The spectra for a preferred embodiment of a Weaver modulator operating to produce vestigial sideband modulation are illustrated in FIGS. 5a-5e. A Weaver modulator modified to produce the spectra of FIGS. 5a-5e processes baseband spectrum 80, centered about DC, i.e., xcfx89=0, as illustrated in FIG. 5a. Spectrum 80 includes negative and positive frequency portions 82 and 84 which are mirror images of each other, so that negative frequency portion 82 extends from xcfx89=0 to xe2x88x92xcfx893, and the positive frequency portion 84 extends from xcfx89=0 to +xcfx893.
Spectra 80 of FIGS. 5a and 5b are multiplied by cosine and sine waves having a frequency xcfx894, which is preferably less than or equal to       ω    3    2
and must be more than 0. In consequence, folded baseband I channel spectra 90 and 92 are derived by multiplying the cosine wave having a frequency xcfx894 (indicated by line 86) by spectrum 80.
Folded baseband Q channel spectra 94 and 96 are derived by multiplying the sine wave having a frequency xcfx894 (indicated by line 88) by spectrum 80. I channel spectra 90 and 92 have phases that are orthogonal to Q channel spectra 94 and 96. The positive frequencies in spectra 90 and 94 respectively represent the folded baseband I and Q channels of spectrum 80 for frequencies below xcfx894, while the positive frequencies in spectra 92 and 96 include the portions of spectrum 80 having frequencies in excess of xcfx894. Spectra 90, 92, 94 and 96 are all centered about xcfx894, with the portions of spectra 90-96 on the left side of xcfx894 being mirror images of the portions of spectra 90-96 on the right side of xcfx894. The sum of spectra 90 and 92 and the sum of spectra 94 and 96 have zero amplitude where they intersect the xcfx89 axis at frequencies +(xcfx893xe2x88x92xcfx894) and xe2x88x92(xcfx893xe2x88x92xcfx894).
Spectra 90 and 92 are applied to a first lowpass filter to derive folded and filtered baseband I spectra 98 and 100, respectively, while spectra 94 and 96 are supplied to a second lowpass filter to derive folded and filtered baseband Q spectra 102 and 104, respectively. The first and second lowpass filters are identical, with each having a cutoff frequency, xcfx895, between (xcfx893+xcfx894) and (xcfx893xe2x88x92xcfx894), so that positive frequency portions of spectra 92 and 96 are essentially unchanged by the lowpass filter, but the higher positive frequency portions of spectra 90 and 94 are severely attenuated, as indicated by steep skirts 106 of spectra 102. Spectra 98-104 are symmetrical with and are mirror images about xcfx89=0.
Spectra 98 and 100 are multiplied by a cosine wave (represented by line 105, FIG. 5c,) at intermediate frequency xcfx89IF, to derive folded and filtered I channel spectra 108 and 110. Spectra 108 and 110, centered at frequency xcfx89IF, are respectively substantial replicas of spectra 98 and 100. Spectra 102 and 104 are multiplied by a sine wave (represented by line 107, FIG. 5d) at frequency xcfx89IF to derive folded and filtered Q-channel spectra 112 and 114, both centered at frequency xcfx89IF and having substantially the same shape as spectra 102 and 104. The phases of spectra 108 and 110 are orthogonal to those of spectra 112 and 114. Spectra 108-114 are linearly combined, i.e., added or subtracted, to produce a vestigial sideband signal.
In FIG. 5e, spectra 108-114 are added to produce vestigial sideband spectrum 116, including relatively steep skirt 118 that is a replica of steep skirts 106 of spectra 98 and 102. Spectrum 116 also includes (1) segment 120 which increases gradually to a peak value from the maximum amplitude of steep skirt 118, and (2) gradually decreasing portion 122 extending from the peak amplitude of spectrum 116 to zero amplitude at the w axis and which includes xcfx89IF. The vestigial sideband includes all of steep skirt 118 and portion 120 of spectrum 116. In ATSC modulation, where a vestige of a replicated inverted spectrum of a discrete time digital signal appears as a tail of an upper sideband (lower sideband at I.F.) full sideband portion 122 also includes a Nyquist frequency, xcfx89N, which is spaced from xcfx89IF by xc2xdxcfx895xe2x88x92xcfx894, where xcfx895 is the sampling frequency.
A comparison of spectra 76 and 116 indicates the portion of spectra 116 that is the vestigial sideband. Vestigial sideband spectrum 116 differs from single sideband spectrum 76 because spectrum 116 contains a vestige of the opposite sideband arising from the mirror image negative frequencies 82 of spectrum 80. In this regard, steep skirt 118 is to the left of carrier frequency xcfx89IFxe2x88x92xcfx894 in spectrum 116; carrier frequency xcfx89IFxe2x88x92xcfx894 in spectrum 116, at the highest amplitude value in the spectrum of FIG. 5E, corresponds to xcfx89cxe2x88x92xcfx89f in spectrum 76. The vestige also includes portion 120 of spectrum 116 that extends to the left of xcfx89IFxe2x88x92xcfx894.
The high frequency end of vestigial sideband spectrum 116 also includes a spectrum tail. In a discrete time system (i.e., a sampled system), the spectrum tail arises from a small portion of the inverted replicated frequency just beyond xcfx89N, the Nyquist frequency.
The amplitude versus frequency response spectrum diagrams of FIGS. 5f and 5g are helpful in understanding the operations described in connection with FIGS. 5a-5e. FIG. 5f is an illustration of a theoretical baseband input spectrum applied to a modified Weaver modulator according to the invention. The input spectrum of FIG. 5f has a constant, non-zero amplitude between angular frequencies xe2x88x92xcfx893 and +W3, and a zero amplitude for |xcfx89| greater than xcfx893. The modified Weaver modulator converts the baseband input of FIG. 5f into the vestigial sideband spectrum of FIG. 5g, having a carrier frequency of xcfx89c; xcfx89c is either an R.F. or I.F. carrier. The spectrum of FIG. 5g has a constant, non-zero amplitude between frequencies (xcfx89cxe2x88x92xcfx891) and (xcfx89c+xcfx893). At xcfx89c+xcfx893, the vestigial sideband spectrum of FIG. 5g has a step drop from the constant, non-zero amplitude to a zero value. The vestigial sideband spectrum has a frequency transition range from (xcfx89cxe2x88x92xcfx891) to (xcfx892xe2x88x92xcfx892) between the constant, non-zero and zero amplitude levels. The angular frequencies xe2x88x92xcfx891 and xe2x88x92xcfx892 in the baseband input of FIG. 5f are in the xe2x80x9cnegativexe2x80x9d frequency portion of the baseband input, such that the absolute value of xcfx891 is less than the absolute value of xcfx892.
To achieve the vestigial sideband spectrum of FIG. 5g, the lowpass filters of the modified Weaver modulator must completely reject all angular frequencies in excess of the sum of the folding frequency (xcfx89f) and xcfx892 and the folding frequency must exceed                     ω        3            -              ω        2              2    .
The lowpass filters must pass, without attenuation, the greater of |xcfx893xe2x88x92xcfx89F| and |xcfx89F+xcfx891|. The lowpass filters must have a response of the type generally indicated by the amplitude versus frequency response curve of FIG. 5h. Each of the lowpass filters has a substantially zero attenuation between DC, where xcfx89=0, and the greater of |xcfx893xe2x88x92xcfx89f| and |xcfx89f+xcfx891|. Each of the lowpass filters completely rejects all frequencies greater than (xcfx89fxe2x88x92xcfx892)
In the case of an 8 or 16 level ATSC digital television signal, filtering of the vestigial sideband signal is symmetrical. Therefore, the shape of the vestigial lower sideband is the same as the shape of the inverted replicated spectrum appearing around the Nyquist rate.
The action of the modified Weaver modulator in combining the positive and negative frequencies of spectrum 80 and the positive and negative replicated frequencies resulting from the multiplying action of the sine and cosine waves at folding frequency xcfx894 is illustrated in FIG. 6 which represents the digital signal derived from multiplier 18 at baseband. In FIG. 6, block 124, indicates the positive frequencies of spectrum 80, extending between 0 and xcfx89m while block 126 indicates the positive replicated frequencies having an inverted spectrum between frequencies xcfx89m and 2xcfx89m. Block 128, extending between xcfx89=0 and xcfx89=xe2x88x92m, indicates the negative frequencies with an inverted spectrum 82 of spectrum 80. The negative replicated frequencies of spectrum 82, between frequencies xe2x88x92m and xe2x88x922xcfx89m, are indicated by block 130.
It is, accordingly, an object of the present invention to provide a new and improved method of and apparatus for generating a vestigial sideband signal, particularly a digital vestigial sideband I.F. signal.
Another object of the invention is to provide a new and improved method of and apparatus for using the Weaver modulation technique for deriving a vestigial sideband signal.
A further object of the invention is to provide a new and improved digital television transmitter apparatus and method.
An added object of the invention is to provide a digital television transmitter for deriving a vestigial sideband signal directly, without filtering a double sideband signal and without using a Hilbert transform.
An additional object of the present invention is to provide a new and improved digital television transmitter including a vestigial sideband generator including a finite impulse response filter having a relatively low number of filter coefficients, while still obtaining a favorable root-raised cosine response.
Yet another object of the invention is to provide a new and improved easily implemented digital television transmitter for deriving a vestigial sideband signal having a favorable root-raised cosine response which does not require a low frequency or high frequency equalizer.
A still further object of the invention is to provide a television transmitter wherein the same digital vestigial sideband arrangement can be used to produce NTSC and ATSC signals.
Still another object of the invention is to provide an ATSC television transmitter for deriving a digital vestigial I.F. signal having a sampling frequency at least twice as high as prior art transmitters and finite impulse response filters having a relatively low number of filter coefficients, while still obtaining a favorable root-raised cosine response.
In accordance with one aspect of the present invention a first signal having a frequency xcfx89m in a predetermined bandwidth extending from xcfx891 to xcfx892 is converted to a vestigial sideband signal by multiplying the first signal by orthogonally phased sinusoidal components having a frequency xcfx894 to derive orthogonal second and third signals each having a frequency equal to (xcfx89mxe2x88x92xcfx894) and containing the information in the first signal. The second and third signals are lowpass filtered to derive fourth and fifth orthogonal signals each having the same frequency and containing the information in the first signal. The frequencies xcfx891, xcfx892 and xcfx894 are such that a vestigial sideband signal modulating the carrier and containing the information in the first signal is derived when (1) the fourth and fifth signals are multiplied with quadrature phases of a carrier and (2) the resulting product signals are linearly combined. Signals that substantially replicate the information in the fourth and fifth signals and a carrier are combined to derive the vestigial sideband signal containing the information in the first signal.
In a preferred embodiment, xcfx894 is a folding frequency greater than zero and less than or equal to                     ω        1            +              ω        2              2    .
The fourth and fifth signals (1) have a spectrum extending between xcfx891 and xcfx893, and (2) are such that the frequency components of the first signal extending between xcfx891 and xcfx894 are folded on the frequency components of the first signal extending between xcfx894 and xcfx892. Signals that substantially replicate the information in the fourth and fifth signals are combined to derive the vestigial sideband signal.
In the preferred embodiments, xcfx894 is a folding frequency no greater than       ω    2    2
and the first signal is a baseband signal such that xcfx891=0. The low pass filtering step passes without substantial attenuation signals having frequencies from DC to xcfx895 and substantially rejects frequencies greater than xcfx896, where xcfx895 is    greater than             ω      2        2  
and xcfx896 is  less than xcfx892. The combining step is such that negative frequencies of the first signal appear in the vestigial sideband signal as a vestige of the sideband of the vestigial sideband signal containing the positive frequencies of the first signal.
The first, second, third, fourth and fifth signals are preferably digital signals having a first fixed sampling frequency. Circuitry responsive to the fourth and fifth signals converts the digital information in the fourth and fifth signals into an analog vestigial sideband signal modulating an R.F. carrier. In one embodiment the fourth and fifth digital signals are converted directly into analog signals that are separately mixed with orthogonal phases of an R.F. carrier and the resulting product signals are linearly combined.
In other, more preferred embodiments, the sampling frequency of the fourth and fifth signals is increased to a second fixed sampling frequency. In some preferred embodiments the signals having the second fixed sampling frequency are mixed with orthogonal phases of a digital I.F. carrier to derive first and second digital product signals that are linearly combined to derive a digital vestigial I.F. In another embodiment, the digital signals at the second sampling frequency are combined with several orthogonal phases of a digital I.F. carrier to derive a pair of orthogonal I.F. digital signals that are converted into analog signals which are, in turn, separately mixed with orthogonal phases of an R.F. carrier. The resulting R.F. modulated waves are linearly combined to derive an R.F. vestigial sideband signal.
In some embodiments the sampling frequency of the fourth and fifth digital signals is offset to a frequency having a round number value. In one embodiment offsetting is achieved by arranging the first signal so it has a sampling frequency that is a fraction of a fixed sampling frequency of an input signal containing the information in the first signal. In such a case, offsetting is achieved by increasing the sampling frequency of the input signal to derive the first signal and low pass filtering the first signal to derive the signal multiplied by the components having the frequency xcfx894. In another embodiment, offsetting is achieved with circuitry including a low frequency digital source deriving signals representing a pair of orthogonal sinusoidal waves. Digital circuitry including multiple multipliers and adders responds to the low pass filtered signals and the signals representing the orthogonal sinusoidal waves to derive signals that are applied to circuits for increasing the sampling frequency.
In a preferred embodiment, the first signal is an ATSC or NTSC digital television signal and the vestigial sideband signal results from at least one digital signal that contains the information necessary to derive the vestigial sideband transmitted signal. In a preferred embodiment the digital signal is an I.F. that is converted to an analog I.F. signal. The I.F. is up frequency converted to derive an I.F. output signal having a spectrum inverted relative to the I.F. signal.
Preferably, for the ATSC signal, each cycle of each of the orthogonally sinusoidal phased components that are multiplied with the input signal is represented by four digital values. In one embodiment, each of the multiplying operations is performed such that a first of the digital values passes bits of each of the multiplied signals in unaltered form and a second of the digital values inverts the polarity of each of the signals. In a second embodiment, each of the multiplying operations is performed such that a first of the digital values passes bits of each of the multiplied signals in unaltered form, a second of the digital inverts the polarity of bits of each of the signals and a third of the digital values blocks bits of each of the multiplied signals.
When the digital television signal is an NTSC signal the frequency modulated aural carrier may be added by adding quadrature FM components at the proper frequency to the folded baseband signals.
Another aspect of the invention relates to a digital television transmitter responsive to a digital television signal. The transmitter includes a digital vestigial sideband modulator including a digital sinusoidal source for deriving a digital signal representing a sinusoidal wave. A digital multiplier multiplies a digital signal including the information in the digital television signal by the digital signal representing the sinusoidal wave: to derive plural digital product signals. A digital lowpass filter arrangement passes low frequency components of the plural digital product signals and blocks high frequency components of the plural digital product signals. Circuitry responsive to a carrier and the signals passed by the low pass filter arrangement derives an analog vestigial sideband signal including the information in the digital television signal.
In one preferred embodiment, the circuitry includes a digital linear combiner for deriving a digital vestigial sideband I.F. signal having a frequency of about 20 mHz. A digital to analog converter arrangement converts the digital vestigial sideband I.F. signal into an analog intermediate frequency signal. An up converter increases the frequency of the intermediate frequency to a desired transmission frequency or channel.